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general description the max1951/MAX1952 high-efficiency, dc-to-dc step-down switching regulators deliver up to 2a of out- put current. the devices operate from an input voltage range of 2.6v to 5.5v and provide an output voltage from 0.8v to v in , making the max1951/MAX1952 ideal for on-board postregulation applications. the max1951 total output error is less than 1% over load, line, and temperature. the max1951/MAX1952 operate at a fixed frequency of 1mhz with an efficiency of up to 94%. the high operating frequency minimizes the size of external components. internal soft-start control circuitry reduces inrush current. short-circuit and thermal-overload protection improve design reliability. the max1951 provides an adjustable output from 0.8v to v in , whereas the MAX1952 has a preset output of 1.8v. both devices are available in a space-saving 8-pin so package. applications asic/dsp/?/fpga core and i/o voltages set-top boxes cellular base stations networking and telecommunications features compact 0.385in 2 circuit footprint 10? ceramic input and output capacitors, 2? inductor for 1.5a output efficiency up to 94% 1% output accuracy over load, line, and temperature (max1951, up to 1.5a) guaranteed 2a output current operate from 2.6v to 5.5v supply adjustable output from 0.8v to v in (max1951) preset output of 1.8v (1.5% accuracy) (MAX1952) internal digital soft-soft short-circuit and thermal-overload protection max1951/MAX1952 1mhz, all-ceramic, 2.6v to 5.5v input, 2a pwm step-down dc-to-dc regulators ________________________________________________________________ maxim integrated products 1 ordering information 19-2622; rev 1; 8/03 for pricing, delivery, and ordering information, please contact maxim/dallas direct! at 1-888-629-4642, or visit maxim? website at www.maxim-ic.com. part temp range pin- package output max1951 esa -40 c to +80 c 8 so adj 0.8v to v in MAX1952 esa -40 c to +80 c 8 so fixed 1.8v max1951 in off on lx comp output 0.8v to v in , up to 2a input 2.6v to 5.5v ref fb pgnd gnd v cc optional typical operating circuit pin configuration pgnd comp fb 1 2 8 7 in lx ref gnd v cc so top view 3 4 6 5 max1951 MAX1952
max1951/MAX1952 1mhz, all-ceramic, 2.6v to 5.5v input, 2a pwm step-down dc-to-dc regulators 2 _______________________________________________________________________________________ absolute maximum ratings electrical characteristics (v in = v cc = 3.3v, pgnd = gnd, fb in regulation, c ref = 0.1?, t a = 0? to +85? , unless otherwise noted. typical values are at t a = +25?.) stresses beyond those listed under ?bsolute maximum ratings?may cause permanent damage to the device. these are stress rating s only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specificatio ns is not implied. exposure to absolute maximum rating conditions for extended periods may affect device reliability. parameter conditions min typ max units in and v cc in voltage range 2.6 5.5 v supply current switching with no load, lx floating v in = 5.5v 6 10 ma shutdown current comp = gnd 0.5 1.0 ma v cc rising 2.35 2.5 v cc undervoltage lockout threshold when lx starts/stops switching v cc falling 2 2.25 v ref ref voltage i ref = 0, v in = 2.6v to 5.5v 1.96 2 2.03 v ref load regulation i ref = 0 to 40a, v in = 2.6v to 5.5v 0.01 0.2 % ref line regulation i ref = 20a, v in = 2.6v to 5.5v 0.01 0.4 % ref shutdown resistance from ref to gnd, comp = gnd 12 22 ? comp max1951 40 60 80 comp transconductance from fb to comp, v comp = 1.25v MAX1952 26.7 40 53.3 s comp clamp voltage, low v in = 2.6v to 5.5v, v fb = 1.3v 0.6 1 1.2 v comp clamp voltage, high v in = 2.6v to 5.5v, v fb = 1.1v 1.97 2.15 2.28 v comp shutdown resistance from comp to gnd, v in = 2v 15 30 ? comp rising 0.6 1 comp shutdown threshold when lx starts/stops switching comp falling 0.17 0.4 v comp startup current comp = gnd 15 25 40 a fb output voltage range (max1951) when using external feedback resistors to drive fb 0.8 v in v v in = 2.6v to 5.5v max1951 0.787 0.795 0.803 fb regulation voltage (error amp only) v comp = 1v to 2v, i out = 0 to 1.5a v in = 2.8v to 5.5v MAX1952 1.773 1.8 1.827 v fb input resistance MAX1952 13 18 28 k ? fb input bias current max1951 -0.1 +0.1 a in, v cc to gnd ........................................................-0.3v to +6v comp, fb, ref to gnd .............................-0.3v to (v cc + 0.3v) lx to current (note 1).........................................................4.5a pgnd to gnd .............................................internally connected continuous power dissipation (t a = +85 c) 8-pin so (derate 12.2mw/ c above +70 c)................976mw operating temperature range max195_ esa..................................................-40 c to +85 c junction temperature range ............................-40 c to +150 c storage temperature range .............................-65 c to +150 c lead temperature (soldering, 10s) .................................+300 c note 1: lx has internal clamp diodes to pgnd and in. applications that forward bias these diodes should take care not to exceed the ic s package power dissipation limits.
max1951/MAX1952 1mhz, all-ceramic, 2.6v to 5.5v input, 2a pwm step-down dc-to-dc regulators _______________________________________________________________________________________ 3 electrical characteristics (continued) (v in = v cc = 3.3v, pgnd = gnd, fb in regulation, c ref = 0.1f, t a = 0 c to +85 c , unless otherwise noted. typical values are at t a = +25 c.) parameter conditions min typ max units lx v in = 5v 116 v in = 3.3v 140 266 lx on-resistance, pmos v in = 2.6v 163 m ? v in = 5v 93 v in = 3.3v 106 206 lx on-resistance, nmos v in = 2.6v 116 m ? lx current-sense transimpedance from lx to comp, v in = 2.6v to 5.5v 0.16 0.24 0.35 ? high side 2.4 3.1 4.5 lx current-limit threshold duty cycle = 100%, v in = 2.6v to 5.5v low side -0.6 a v lx = 5.5v 10 lx leakage current v in = 5.5v lx = gnd -10 a lx switching frequency v in = 2.6v to 5.5v 0.85 1 1.1 mhz lx maximum duty cycle v comp = 1.5v, lx = high-z, v in = 2.6v to 5.5v 100 % lx minimum duty cycle v comp = 1v, v in = 2.6v to 5.5v 15 % thermal characteristics t j rising 160 thermal-shutdown threshold when lx starts/stops switching t j falling 145 c electrical characteristics (v in = v cc = 3.3v, pgnd = gnd, fb in regulation, c ref = 0.1f, t a = -40 c to +85 c , unless otherwise noted.) (note 2) parameter conditions min typ max units in and v cc in voltage range 2.6 5.5 v supply current switching with no load, v in = 5.5v 10 ma shutdown current comp = gnd 1 ma v cc rising 2.5 v cc undervoltage lockout threshold when lx starts/stops switching v cc falling 1.95 v ref ref voltage i ref = 0, v in = 2.6v to 5.5v 1.95 2.03 v ref load regulation i ref = 0 to 40a, v in = 2.6v to 5.5v 0.2 % ref line regulation i ref = 20a, v in = 2.6v to 5.5v 0.4 % ref shutdown resistance from ref to gnd, comp = gnd 22 ?
max1951/MAX1952 1mhz, all-ceramic, 2.6v to 5.5v input, 2a pwm step-down dc-to-dc regulators 4 _______________________________________________________________________________________ note 2: specifications to -40 c are guaranteed by design and not production tested. note 3: the lx output is designed to provide 2.4a rms current. electrical characteristics (continued) (v in = v cc = 3.3v, pgnd = gnd, fb in regulation, c ref = 0.1f, t a = -40 c to +85 c , unless otherwise noted.) (note 2) parameter conditions min typ max units comp max1951 40 80 comp transconductance from fb to comp, v comp = 1.25v MAX1952 26.7 53.3 s comp clamp voltage, low v in = 2.6v to 5.5v, v fb = 1.3v 0.6 1.2 v comp clamp voltage, high v in = 2.6v to 5.5v, v fb = 1.1v 1.97 2.28 v comp shutdown resistance from comp to gnd, v in = 2v 30 ? comp rising 1.2 comp shutdown threshold when lx starts/stops switching comp falling 0.17 v comp startup current comp = gnd 14 40 a fb output voltage range (max1951) when using external feedback resistors to drive fb 0.8 v in v max1951 0.783 0.807 fb regulation voltage (error amp only) v comp = 1v to 2v, v in = 2.6v to 5.5v MAX1952 1.764 1.836 v fb input resistance from fb to gnd MAX1952 10 30 k ? fb input bias current max1951 -0.1 +0.1 a lx lx on-resistance, pmos 266 m ? lx on-resistance, nmos 206 m ? lx current sense from lx to comp, v in = 2.6v to 5.5v 0.16 0.35 ? lx current-limit threshold duty cycle = 100%, v in = 2.6v to 5.5v, high side 2.4 4.5 a v lx = 5.5v 10 lx leakage current v in = 5.5v lx = gnd -10 a lx switching frequency v in = 2.6v to 5.5v 0.8 1.1 mhz lx maximum duty cycle v comp = 1.5v, lx = hi-z, v in = 2.6v to 5.5v 100 %
max1951/MAX1952 1mhz, all-ceramic, 2.6v to 5.5v input, 2a pwm step-down dc-to-dc regulators _______________________________________________________________________________________ 5 efficiency vs. load current (v cc = v in = 5v) max 1951 toc01 load current (ma) efficiency (%) 1000 100 10 20 30 40 50 60 70 80 90 100 0 10 10,000 v out = 3.3v v out = 2.5v v out = 1.5v v out = 0.8v efficiency vs. load current (v cc = v in = 3.3v) max 1951 toc02 load current (ma) efficiency (%) 1000 100 10 20 30 40 50 60 70 80 90 100 0 10 10,000 v out = 2.5v v out = 1.8v v out = 1.5v v out = 0.8v ref voltage vs. ref output current max1951 toc03 ref output current ( a) ref voltage (v) 35 30 25 20 15 10 5 1.990 1.991 1.992 1.993 1.994 1.995 1.989 040 t a = +85 c t a = +25 c t a = -40 c switching frequency vs. input voltage max1951 toc04 input voltage (v) switching frequency (mhz) 5.1 4.6 3.1 3.6 4.1 0.85 0.90 0.95 1.00 1.05 1.10 1.15 1.20 0.80 2.6 5.6 t a = +85 c t a = +25 c t a = -40 c output voltage deviation vs. load current max1951 toc05 load current (a) output voltage deviation (mv) 1.2 0.8 0.4 -5 -4 -3 -2 -1 0 1 2 3 4 5 6 -6 0 1.6 v out = 2.5v v out = 3.3v v out = 0.8v v out = 1.8v typical operating characteristics (typical values are at v in = v cc = 5v, v out = 1.5v, i out = 1.5a, and t a = +25 c, unless otherwise noted. see figure 2.) load transient response max1951 toc06 40 s/div 0 output voltage: 100mv/div, ac-coupled output current: 0.5a/div v in = 5v v out = 2.5v i out = 0.5 to 1a load transient response max1951 toc07 40 s/div 0 output voltage: 100mv/div, ac-coupled output current: 0.5a/div v in = 3.3v v out = 1.5v i out = 0.5 to 1a
max1951/MAX1952 1mhz, all ceramic, 2.6v to 5.5v input, 2a pwm step-down dc-to-dc regulators 6 _______________________________________________________________________________________ typical operating characteristics (continued) (typical values are at v in = v cc = 5v, v out = 1.5v, i out = 1.5a, and t a = +25 c, unless otherwise noted. see figure 2.) shutdown current vs. input voltage max1951 toc12 input voltage (v) shutdown current (ma) 5.0 4.5 4.0 3.5 3.0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 0 2.5 5.5 switching waveforms max1951 toc08 200ns/div 0 0 inductor current 1a/div v lx 5v/div output voltage 10mv/div, ac-coupled v in = 3.3v v out = 1.8v i load = 1.5a soft-start waveforms max1951 toc09 1ms/div v comp 2v/div output voltage 1v/div v in = v cc = 3.3v v out = 2.5v i load = 1.5a soft-start waveforms max1951 toc10 1ms/div v comp 2v/div output voltage 0.5v/div v in = v cc = 3.3v v out = 0.8v shutdown waveforms max1951 toc11 20 s/div 0 0 0 v comp 2v/div v lx 5v/div output voltage 1v/div v in = v cc = 3.3v v out = 2.5v i load = 1.5a
max1951/MAX1952 1mhz, all-ceramic, 2.6v to 5.5v input, 2a pwm step-down dc-to-dc regulators _______________________________________________________________________________________ 7 detailed description the max1951/MAX1952 high-efficiency switching regula- tors are small, simple, dc-to-dc step-down converters capable of delivering up to 2a of output current. the devices operate in pulse-width modulation (pwm) at a fixed frequency of 1mhz from a 2.6v to 5.5v input voltage and provide an output voltage from 0.8v to v in , making the max1951/MAX1952 ideal for on-board postregula- tion applications. the high switching frequency allows for the use of smaller external components, and internal synchronous rectifiers improve efficiency and eliminate the typical schottky free-wheeling diode. using the on- resistance of the internal high-side mosfet to sense switching currents eliminates current-sense resistors, further improving efficiency and cost. the max1951 total output error over load, line, and temperature (0 c to +85 c) is less than 1%. controller block function the max1951/MAX1952 step-down converters use a pwm current-mode control scheme. an open-loop com- parator compares the integrated voltage-feedback signal against the sum of the amplified current-sense signal and the slope compensation ramp. at each rising edge of the internal clock, the internal high-side mosfet turns on until the pwm comparator trips. during this on-time, cur- rent ramps up through the inductor, sourcing current to the output and storing energy in the inductor. the current- mode feedback system regulates the peak inductor cur- rent as a function of the output voltage error signal. since the average inductor current is nearly the same as the peak inductor current (<30% ripple current), the circuit acts as a switch-mode transconductance amplifier. to preserve inner-loop stability and eliminate inductor stair- casing, a slope-compensation ramp is summed into the main pwm comparator. during the second half of the cycle, the internal high-side p-channel mosfet turns off, and the internal low-side n-channel mosfet turns on. the inductor releases the stored energy as its current ramps down while still providing current to the output. the output capacitor stores charge when the inductor current exceeds the load current, and discharges when the inductor current is lower, smoothing the voltage across the load. under overload conditions, when the inductor current exceeds the current limit (see the current limit section), the high-side mosfet does not turn on at the rising edge of the clock and the low-side mosfet remains on to let the inductor current ramp down. current sense an internal current-sense amplifier produces a current signal proportional to the voltage generated by the high-side mosfet on-resistance and the inductor cur- rent (r ds(on) x i lx ). the amplified current-sense signal and the internal slope compensation signal are summed together into the comparator s inverting input. the pwm comparator turns off the internal high-side mosfet when this sum exceeds the output from the voltage-error amplifier. current limit the internal high-side mosfet has a current limit of 3.1a (typ). if the current flowing out of lx exceeds this limit, the high-side mosfet turns off and the synchronous rectifier turns on. this lowers the duty cycle and causes the output voltage to droop until the current limit is no longer exceeded. a synchronous rectifier current limit of -0.6a (typ) protects the device from current flowing into lx. if the negative current limit is exceeded, the synchro- nous rectifier turns off, forcing the inductor current to flow pin description pin name function 1v cc supply voltage. bypass with 0.1f capacitor to ground and 10 ? resistor to in. 2 ref reference bypass. bypass with 0.1f capacitor to ground. 3 gnd ground 4fb feedback input. connect to the output to regulate using the internal feedback resistor string (MAX1952). connect an external resistor- divider from the output to fb and gnd to set the output to a voltage between 0.8v and v in (max1951). 5 comp regulator compensation. connect series rc network to gnd. pull comp below 0.17v to shut down the regulator. comp = gnd when v in is less than 2.25v (see the compensation and shutdown mode section) 6 pgnd power ground. internally connected to gnd. keep power ground and signal ground planes separate. 7lx inductor connection. connect an inductor between lx and the regulator output. 8in power-supply voltage. input voltage range from 2.6v to 5.5v. bypass with a 10f (min) ceramic capacitor to gnd and a 10 ? resistor to v cc .
max1951/MAX1952 1mhz, all-ceramic, 2.6v to 5.5v input, 2a pwm step-down dc-to-dc regulators 8 _______________________________________________________________________________________ through the high-side mosfet body diode, back to the input, until the beginning of the next cycle or until the inductor current drops to zero. the max1951/MAX1952 utilize a pulse-skip mode to prevent overheating during short-circuit output conditions. the device enters pulse- skip mode when the fb voltage drops below 300mv, lim- iting the current to 3a (typ) and reducing power dissipation. normal operation resumes upon removal of the short-circuit condition. v cc decoupling due to the high switching frequency and tight output tolerance (1%), decouple v cc with a 0.1f capacitor connected from v cc to gnd, and a 10 ? resistor con- nected from v cc to in. place the capacitor as close to v cc as possible. soft-start the max1951/MAX1952 employ digital soft-start circuitry to reduce supply inrush current during startup conditions. when the device exits undervoltage lockout (uvlo), shut- down mode, or restarts following a thermal-overload event, or the external pulldown on comp is released, the digital soft-start circuitry slowly ramps up the voltages at ref and fb (see the soft-start waveforms in the typical operating characteristics) . undervoltage lockout if v cc drops below 2.25v, the uvlo circuit inhibits switching. once v cc rises above 2.35v, the uvlo clears, and the soft-start sequence activates. compensation and shutdown mode the output of the internal transconductance voltage error amplifier connects to comp. the normal opera tion voltage for comp is 1v to 2.2v. to shut down the max1951/MAX1952, use an npn bipolar junction transistor or a very low output capacitance open-drain mosfet to pull comp to gnd. shutdown mode causes the internal mosfets to stop switching, forces lx to a high-impedance state, and shorts ref to gnd. release comp to exit shutdown and initiate the soft- start sequence. thermal-overload protection thermal-overload protection limits total power dissipation in the device. when the junction temperature exceeds t j = +160 c, a thermal sensor forces the device into shut- down, allowing the die to cool. the thermal sensor turns the device on again after the junction temperature cools by 15 c, resulting in a pulsed output during continuous overload conditions. following a thermal-shutdown condi- tion, the soft-start sequence begins. max1951 v cc comp current sense slope comp error signal clock positive and negative current limits gnd ref fb pgnd lx in dac g m soft-start/ uvlo ref 2v pwm control thermal shutdown bandgap ref 1.25v clamp osc ramp gen figure 1. functional diagram
max1951/MAX1952 1mhz, all-ceramic, 2.6v to 5.5v input, 2a pwm step-down dc-to-dc regulators _______________________________________________________________________________________ 9 design procedure output voltage selection: adjustable (max1951) or preset (MAX1952) the max1951 provides an adjustable output voltage between 0.8v and v in . connect fb to output for 0.8v output. to set the output voltage of the max1951 to a voltage greater than v fb (0.8v typ), connect the output to fb and gnd using a resistive divider, as shown in figure 2a. choose r2 between 2k ? and 20k ? , and set r3 according to the following equation: r3 = r2 x [(v out / v fb ) 1] the max1951 pwm circuitry is capable of a stable min- imum duty cycle of 18%. this limits the minimum output voltage that can be generated to 0.18 ? v in . instability may result for v in /v out ratios below 0.18. the MAX1952 provides a preset output voltage. connect the output to fb, as shown in figure 2b. output inductor design use a 2h inductor with a minimum 2a-rated dc cur- rent for most applications. for best efficiency, use an inductor with a dc resistance of less than 20m ? and a saturation current greater than 3a (min). see table 2 for recommended inductors and manufacturers. for most designs, derive a reasonable inductor value (l init ) from the following equation: l init = v out x (v in - v out ) / (v in x lir x i out(max) x f sw ) where f sw is the switching frequency (1mhz typ) of the oscillator. keep the inductor current ripple percentage lir between 20% and 40% of the maximum load cur- rent for the best compromise of cost, size, and perfor- mance. calculate the maximum inductor current as: i l(max) = (1 + lir / 2) x i out(max) check the final values of the inductor with the output ripple voltage requirement. the output ripple voltage is given by: v ripple = v out x (v in - v out ) x esr / (v in x l final x f sw ) where esr is the equivalent series resistance of the output capacitors. input capacitor design the input filter capacitor reduces peak currents drawn from the power source and reduces noise and voltage ripple on the input caused by the circuit s switching. the input capacitor must meet the ripple current requirement (i rms ) imposed by the switching currents defined by the following equation: for duty ratios less than 0.5, the input capacitor rms current is higher than the calculated current. therefore, use a +20% margin when calculating the rms current at lower duty cycles. use ceramic capacitors for their low esr, equivalent series inductance (esl), and lower cost. choose a capacitor that exhibits less than 10 c temperature rise at the maximum operating rms cur- rent for optimum long-term reliability. after determining the input capacitor, check the input ripple voltage due to capacitor discharge when the high-side mosfet turns on. calculate the input ripple voltage as follows: v in_ripple = (i out x v out ) / (f sw x v in x c in ) keep the input ripple voltage less than 3% of the input voltage. output capacitor design the key selection parameters for the output capacitor are capacitance, esr, esl, and the voltage rating requirements. these affect the overall stability, output ripple voltage, and transient response of the dc-to-dc converter. the output ripple occurs due to variations in the charge stored in the output capacitor, the voltage drop due to the capacitor s esr, and the voltage drop due to the capacitor s esl. calculate the output voltage ripple due to the output capacitance, esr, and esl as: v ripple = v ripple(c) + v ripple(esr) + v ripple(esl) where the output ripple due to output capacitance, esr, and esl is: v ripple(c) = i p-p / (8 x c out x f sw ) v ripple(esr) = i p-p x esr v ripple(esl) = (i p-p / t on ) x esl or (i p-p / t off ) x esl, whichever is greater and i p-p the peak-to-peak inductor current is: i p-p = [ (v in v out ) / f sw x l) ] x v out / v in use these equations for initial capacitor selection, but determine final values by testing a prototype or evalua- tion circuit. as a rule, a smaller ripple current results in less output voltage ripple. since the inductor ripple current is a factor of the inductor value, the output voltage ripple decreases with larger inductance. use ceramic capacitors for their low esr and esl at the switching frequency of the converter. the low esl of ceramic capacitors makes ripple voltages negligible. load transient response depends on the selected output capacitor. during a load transient, the output instantly changes by esr x i load . before the controller can respond, the output deviates further, depending on the inductor and output capacitor values. after a short time (see the load transient response graph in the ivivvv rms in out out in out = ? (/ ) ( ( )) 1 2
max1951/MAX1952 1mhz, all-ceramic, 2.6v to 5.5v input, 2a pwm step-down dc-to-dc regulators 10 ______________________________________________________________________________________ typical operating characteristic s), the controller responds by regulating the output voltage back to its nominal state. the controller response time depends on the closed-loop bandwidth. a higher bandwidth yields a faster response time, thus preventing the output from deviating further from its regulating value. compensation design the double pole formed by the inductor and output capacitor of most voltage-mode controllers introduces a large phase shift, which requires an elaborate compensa- tion network to stabilize the control loop. the max1951/ MAX1952 utilize a current-mode control scheme that reg- ulates the output voltage by forcing the required current through the external inductor, eliminating the double pole caused by the inductor and output capacitor, and greatly simplifying the compensation network. a simple type 1 compensation with single compensation resistor (r 1 ) and compensation capacitor (c 2 ) creates a stable and high- bandwidth loop. an internal transconductance error amplifier compen- sates the control loop. connect a series resistor and capacitor between comp (the output of the error ampli- fier) and gnd to form a pole-zero pair. the external inductor, internal current-sensing circuitry, output capacitor, and the external compensation circuit deter- mine the loop system stability. choose the inductor and output capacitor based on performance, size, and cost. additionally, select the compensation resistor and capacitor to optimize control-loop stability. the compo- nent values shown in the typical application circuit (figure 2) yield stable operation over a broad range of input-to-output voltages. the basic regulator loop consists of a power modulator, an output feedback divider, and an error amplifier. the power modulator has dc gain set by gmc x r load , with a pole-zero pair set by r load , the output capaci- tor (c out ), and its esr. the following equations define the power modulator: modulator gain: g mod = ? v out / ? v comp = gmc x r load modulator pole frequency: fp mod = 1 / (2 x x c out x (r load +esr)) modulator zero frequency: fz esr = 1 / (2 x x c out x esr) where, r load = v out / i out(max) , and gmc = 4.2s. the feedback divider has a gain of g fb = v fb / v out , where v fb is equal to 0.8v. the transconductance error amplifier has a dc gain, g ea(dc), of 70db. the com- pensation capacitor, c 2, and the output resistance of the error amplifier, r oea (20m ? ), set the dominant pole. c 2 and r 1 set a compensation zero. calculate the dominant pole frequency as: fp ea = 1 / (2 x c c x r oea ) determine the compensation zero frequency is: fz ea = 1 / (2 x c c x r c ) for best stability and response performance, set the closed-loop unity-gain frequency much higher than the modulator pole frequency. in addition, set the closed- loop crossover unity-gain frequency less than, or equal to, 1/5 of the switching frequency. however, set the maximum zero crossing frequency to less than 1/3 of the zero frequency set by the output capacitance and its esr when using poscap, spcap, oscon, or other electrolytic capacitors.the loop-gain equation at the unity-gain frequency is: g ea(fc) x g mod(fc) x v fb / v out = 1 where g ea(fc ) = gm ea x r 1 , and g mod(fc) = gmc x r load x fp mod /f c, where gm ea = 60s . r 1 calculated as: r 1 = v out x k / (gm ea x v fb x g mod(fc) ) where k is the correction factor due to the extra phase introduced by the current loop at high frequencies (>100khz). k is related to the value of the output capacitance (see table 1 for values of k vs. c). set the error-amplifier compensation zero formed by r 1 and c 2 at the modulator pole frequency at maximum load. c 2 is calculated as follows: c 2 = (2 x v out x c out / (r 1 x i out(max) ) as the load current decreases, the modulator pole also decreases; however, the modulator gain increases accordingly, resulting in a constant closed-loop unity- gain frequency. use the following numerical example to calculate r 1 and c 2 values of the typical application circuit of figure 2a. table 1. k value v out = 1.5v i out(max) = 1.5a c out = 10f r esr = 0.010 ? gm ea = 60s description c out (f) k 10 0.55 22 0.47 v al ues ar e for outp ut i nd uctance fr om 1.2h to 2.2h . d o not use outp ut i nd uctor s l ar g er than 2.2h . u se f c = 200kh z to cal cul ate r 1 .
max1951/MAX1952 1mhz, all-ceramic, 2.6v to 5.5v input, 2a pwm step-down dc-to-dc regulators ______________________________________________________________________________________ 11 gmc = 4.2s f switch = 1mhz r load = v out / i out(max) = 1.5v / 1.5 a = 1 ? fp mod = [1 / (2 x c out x (r load + r esr )] = [1 / (2 x 10 10 -6 x (1 + 0.01)] = 15.76khz. fz esr = [1/(2 xc out r esr )] = [1 / (2 x 10 10 -6 0.01)] = 1.59mhz. for 2h output inductor, pick the closed-loop unity-gain crossover frequency (f c ) at 200khz. determine the power modulator gain at f c : g mod(fc ) = gmc r load fp mod / f c = 4.2 1 15.76khz / 200khz = 0.33 then: r 1 = v o x k / (gm ea x v fb x g mod(fc ) ) = (1.5 x 0.55) / (60 10 -6 0.8 0.33) 51.1k ? (1%) c 2 = (2 x v out c out ) / (r c i out(max) ) = (2 1.25 10 10 -6 ) / (51.1k 1.5) 209pf, choose 220pf, 10% applications information pc board layout considerations careful pc board layout is critical to achieve clean and stable operation. the switching power stage requires particular attention. follow these guidelines for good pc board layout: 1) place decoupling capacitors as close to the ic as possible. keep power ground plane (connected to pgnd) and signal ground plane (connected to gnd) separate. 2) connect input and output capacitors to the power ground plane; connect all other capacitors to the signal ground plane. 3) keep the high-current paths as short and wide as possible. keep the path of switching current (c1 to in and c1 to pgnd) short. avoid vias in the switching paths. 4) if possible, connect in, lx, and pgnd separately to a large copper area to help cool the ic to further improve efficiency and long-term reliability. 5) ensure all feedback connections are short and direct. place the feedback resistors as close to the ic as possible. 6) route high-speed switching nodes away from sensi- tive analog areas (fb, comp). thermal considerations the max1951 uses a fused-lead 8-pin so package with a r thjc rating of 32 c/w. the max1951 ev kit layout is optimized for 1.5a. the typical application circuit shown in figure 2c was tested with the existing max1951 ev kit layout at +85 c ambient temperature, and gnd lead temperature was measured at +113 c for a typical device. the estimated junction temperature was +138 c. thermal performance can be further improved with one of the following options: 1) increase the copper areas connected to gnd, lx, and in. 2) provide thermal vias next to gnd and in, to the ground plane and power plane on the back side of pc board, with openings in the solder mask next to the vias to provide better thermal conduction. 3) provide forced-air cooling to further reduce case temperature.
max1951/MAX1952 1mhz, all-ceramic, 2.6v to 5.5v input, 2a pwm step-down dc-to-dc regulators 12 ______________________________________________________________________________________ max1951esa in off r1 51.1k ? r4 10 ? r5 10k ? c5 0.1 f c2 220pf c3 0.1 f r2 16.9k ? 1% r3 14.7k ? 1% l1 2 h c4 10 f c1 10 f on lx comp 1.5v at 1.5a 2.6v to 5.5v ref fb gnd q1 pgnd gnd v cc optional shutdown control component values output voltage (v) 0.8 1.5 2.5 3.3 r1 (k ? ) 33.2 51.1 82.5 110 r2 (k ? ) open 16.9 14 24 r3 (k ? ) short 14.7 30 75 c2 (pf) 220 220 220 220 figure 2a. max1951 adjustable output typical application circuit MAX1952esa-18 in off r1 68k ? r4 10 ? r5 10k ? c5 0.1 f c2 220pf c3 0.1 f l1 2 h c4 10 f c1 10 f on lx comp 1.8v at 1.5a 2.6v to 5.5v ref fb pgnd q1 gnd gnd v cc optional shutdown control figure 2b. MAX1952 fixed-output typical application circuit
max1951/MAX1952 1mhz, all-ceramic, 2.6v to 5.5v input, 2a pwm step-down dc-to-dc regulators ______________________________________________________________________________________ 13 max1951esa in off r1 100k ? r4 10 ? r5 10k ? c5 0.1 f c2 100pf c3 0.1 f l1 1.1 h c4 22 f c1 10 f on lx comp 1.8v, 2a 3.3v 5% ref fb gnd q1 pgnd gnd v cc optional shutdown control l1: toko a915ay-1r1m c1: taiyo yuden jmk316bj106ml c4: taiyo yuden jmk325bj226mm r2 10k ? 1% r3 12.7k ? 1% figure 2c. max1951 typical application circuit with 2a output
max1951/MAX1952 1mhz, all-ceramic, 2.6v to 5.5v input, 2a pwm step-down dc-to-dc regulators 14 ______________________________________________________________________________________ table 2. external components list c o m po n en t ( f i g u r e 2 ) function description l1 output inductor 2h 20% inductor sumida cdrh4d28-1r8 or toko a915ay-2r0m c1 input filtering capacitor 10f 20%, 6.3v x5r capacitor taiyo yuden jmk316bj106ml or tdk c3216x5r0j106mt c2 compensation capacitor 220pf 10%, 50v capacitor murata grm39x7r221k050ad or taiyo yuden umk107ch221kz c3 reference bypass capacitor 0.1f 20%, 16v x7r capacitor taiyo yuden emk107bj104ma, tdk c1608x7r1c104k, or murata grm 39x7r104k016ad c4 output filtering capacitor 10f 20%, 6.3v x5r capacitor taiyo yuden jmk316bj106ml or tdk c3216x5r0j106mt c5 v cc bypass capacitor 0.1f 20%, 16v x7r capacitor taiyo yuden emk107bj104ma, tdk c1608x7r1c104k, or murata grm 39x7r104k016ad r1 loop compensation resistor figure 2a r2 feedback resistor figure 2a r3 feedback resistor figure 2a r4 bypass resistor 10 ? 5% resistor r5 s hutd ow n tr ansi stor b ase cur r ent b i as ( op ti onal ) 10k ? 5% resistor q1 shutdown transistor (optional) npn bipolar junction transistor fairchild mmbt3904 zetex fmmt413 table 3. component suppliers manufacturer phone fax murata 650-964-6321 650-964-8165 sumida 847-545-6700 847-545-6720 taiyo yuden 800-348-2496 847-925-0899 tdk 847-803-6100 847-803-6296 toko 1-800-pik-toko 408-943-9790 chip information transistor count: 2500 process: bicmos
max1951/MAX1952 1mhz, all-ceramic, 2.6v to 5.5v input, 2a pwm step-down dc-to-dc regulators maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a maxim product. no circuit patent licenses are implied. maxim reserves the right to change the circuitry and specifications without notice at any time. maxim integrated products, 120 san gabriel drive, sunnyvale, ca 94086 408-737-7600 ____________________ 15 ? 2003 maxim integrated products printed usa is a registered trademark of maxim integrated products. package information (the package drawing(s) in this data sheet may not reflect the most current specifications. for the latest package outline info rmation go to www.maxim-ic.com/packages .) soicn .eps package outline, .150" soic 1 1 21-0041 b rev. document control no. approval proprietary information title: top view front view max 0.010 0.069 0.019 0.157 0.010 inches 0.150 0.007 e c dim 0.014 0.004 b a1 min 0.053 a 0.19 3.80 4.00 0.25 millimeters 0.10 0.35 1.35 min 0.49 0.25 max 1.75 0.050 0.016 l 0.40 1.27 0.394 0.386 d d mindim d inches max 9.80 10.00 millimeters min max 16 ac 0.337 0.344 ab 8.75 8.55 14 0.189 0.197 aa 5.004.80 8 n ms012 n side view h 0.2440.228 5.80 6.20 e 0.050 bsc 1.27 bsc c h e e b a1 a d 0-8 l 1 variations:

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